High-speed signaling systems and methods with adaptable, continuous-time equalization

ABSTRACT

A receiver includes a continuous-time equalizer, a decision-feedback equalizer (DFE), data and error sampling logic, and an adaptation engine. The receiver corrects for inter-symbol interference (ISI) associated with the most recent data symbol (first post cursor ISI) by establishing appropriate equalization settings for the continuous-time equalizer based upon a measure of the first-post-cursor ISI.

FIELD OF THE INVENTION

The present invention relates generally to the field of communications,and more particularly to high speed electronic signaling within andbetween integrated circuit devices.

BACKGROUND

The performance of many digital systems is limited by theinterconnection bandwidth within and between integrated circuit devices(ICs). High performance communication channels between ICs suffer frommany effects that degrade signals. Primary among them is inter-symbolinterference (ISI) from high frequency signal attenuation andreflections due to impedance discontinuities.

ISI becomes more pronounced at higher signaling rates, ultimatelydegrading signal quality to the point at which distinctions betweenoriginally transmitted signal levels may be lost. Some receivers cancelISI using a decision-feedback equalizer (DFE). DFEs multiply each of Nrecently received symbols by respective tap coefficients, the resultingproducts representing the ISI attributable to the corresponding symbol.The sum of these products is subtracted from the received signal priorto sampling. The ISI associated with the prior data is thereby reducedor eliminated.

In very high-speed systems it can be difficult to resolve the mostrecent data bit or bits in time to calculate their impact on theincoming symbol. Some receivers therefore ignore the impact of suchsymbols on the incoming signal, and consequently fail to correct for theISI attributed to those symbols. Other receivers employ partial responseDFEs (PrDFEs) that obtain multiple samples of the incoming data usingmultiple correction coefficients, one for each of the possible values ofthe most recently received symbol or symbols. The correct sample is thenselected after the most recently received symbol or symbols areresolved.

PrDFEs are effective, but require a separate subtraction and samplingpath for each possible value of the most recently received symbol or, inthe case of multiple symbols (multi-symbol PrDFE), a separatecomputational path for each possible combination of the multiple symbolvalues. This results in e.g. 2^(M) paths in a binary PrDFE system thatconsiders M prior symbols. The additional paths occupy area, requirepower, and slow signal rates by increasing the input capacitance of thereceiver. There is therefore a need for power and area-efficientreceivers capable of filtering incoming signals to cancel ISI from themost recently received symbol or symbols.

BRIEF DESCRIPTION OF THE FIGURES

FIG. 1 depicts a receiver 100, in accordance with one embodiment, thatreceives information from a transmitter (not shown) via a high-speedcommunication channel 105.

FIG. 2 depicts adaptation engine 135 in accordance with one embodiment.

FIG. 3 details an embodiment of a tap-value generator 205 of FIG. 2 thatgenerates a tap value using a sign-sign, least-mean-squared (LMS)algorithm.

FIGS. 4A-4D are waveform diagrams illustrating how tap-value generator205 generates the values for taps α₀ (AGCadj) and α₁ (EQadj) inaccordance with one embodiment.

FIG. 5 depicts three eye diagrams 500, 505, and 510 that illustrate theimpact on an incoming signal Veq′ of adjusting signals AGCadj and EQadj.

FIG. 6 schematically depicts equalizer 120 of FIG. 1 in accordance withone embodiment.

FIG. 7 details an embodiment of variable capacitor 645 of FIG. 6.

FIG. 8 schematically depicts a bias-voltage generator 800 for use withequalizer 120 of FIG. 6.

DETAILED DESCRIPTION

FIG. 1 depicts a receiver 100, in accordance with one embodiment, thatreceives information from a transmitter (not shown) via a high-speedcommunication channel 105. In one embodiment, receiver 100 isinstantiated on an integrated-circuit (IC) device and channel 105provides differential signals RN and RP to a like-named differentialinput port of receiver 100 via a pair of pads 110. Channel 105 is ACcoupled and includes a termination element 115 in this example. In otherembodiments channel 105 is e.g. DC coupled, single ended, or optical. Inembodiments adapted to communicate over optical channels, receiver 100may include an integrated optical-to-electrical converter. Receiver 100includes an analog, continuous-time equalizer 120, a decision-feedbackequalizer (DFE) 125, data and error sampling logic 130, and anequalization-adaptation engine 135.

Equalizer 120 equalizes differential data signal RP/RN, conveyed fromchannel 105 to an input port of equalizer 120, to produce an equalizedsignal Veq on a like-named output port. (As with other designationsherein, Veq refers both to a signal and a corresponding node or port;whether a given designation refers to a signal or a circuit element willbe clear from the context.) Receiver 100 corrects for inter-symbolinterference (ISI) associated with the most recent data symbol (firstpost cursor ISI) by establishing appropriate equalization settings forcontinuous-time equalizer 120 based upon a measure of thefirst-post-cursor ISI. In doing so, receiver 100 can eliminate the needto resolve the most recent data bit in time to calculate its impact onthe incoming signal, and thus facilitate communication at higher speedswithout the attendant complexity and power required by PrDFE-basedreceivers. Some embodiments may use PrDFE for subsequent filter taps orto complement the continuous-time equalizer.

Equalizer 120 amplifies signal RP/RN using a range of amplificationfactors, with higher frequency components typically being treated tohigher amplification factors. Channel 105 will typically exhibit a lowpass filter effect, in which case equalizer 120 may be used tocompensate for attenuation of higher-frequency signal components. Insome embodiments, the low-frequency gain of equalizer 120 may also beadjusted to compensate for broadband signal attenuation. Gainadjustments can be accomplished by frequency-selective amplification orattenuation, or a combination of amplification and attenuation. Ingeneral, the goal of equalization is to reduce or minimize the effectsof ISI, so equalization is typically accomplished by adjusting one ormore characteristics of a signal in a manner that mitigates the effectsof ISI.

DFE 125 further equalizes signal Veq to produce a second equalizedsignal Veq′ for sampling logic 130. DFE 125 stores sequences of sampleddata in a buffer 160 as post-tap data values. Though not shown, tapselect logic may be included to enable selection of a subset of datavalues within buffer 160. Receive-side equalization taps can thus beselected to have latencies that match whatever ISI components areevident in channel 105. Each stored data value in buffer 160 after theinitial latch is multiplied by a corresponding tap coefficient. Theresulting products are summed and the total added to equalized signalVeq to produce the second equalized signal Veq′. In one embodiment clocksignal DfeClk to DFE 125 is a recovered clock signal synchronized to theedges of the equalized signal as observed at the input of sampler 155.The DfeClk is phase offset from (e.g. the complement of) receive clockRClk. The error sampler can be timed to the edges of the equalizedsignal in other embodiments, as by tying the clock terminal of sampler150 to an edge clock signal (not shown).

Amplifier 140 within sampling logic 130 compares signal Veq′ with aselected data level Dlev, outputting a signal indicative of a logic one(zero) if Veq′ is greater than (less than) level Dlev. Sampler 150periodically captures the output from amplifier 140 on rising edges of areceive clock signal RClk to produce a series of error samples Err_(n).A second amplifier 145 compares signal Veq′ with a reference voltage Vr(e.g., zero volts), outputting a signal indicative of a logic one (zero)if Veq′ is greater than (less than) level Vr. Sampler 155 periodicallycaptures the output from amplifier 145 on rising edges of receive clocksignal RClk to produce a series of data samples Data_(n).

Adaptation engine 135 employs data and error samples Data_(n) andErr_(n) from sampling logic 130 to generate the tap values for equalizer120 and DFE 125. In an embodiment in which equalizer 120 is adapted toprovide both automatic gain control (AGC) to compensate for broadbandgain and equalization to compensate for ISI, adaptation engine 135generates measures of DC attenuation and one or more ISI values bycomparing error signals Err_(n) with data samples of various symbollatencies. Based upon these generated values, adaptation engine 135issues low-frequency control signals LFadj and high-frequency controlsignals HFadj to a control port of equalizer 120, and thereby controlsthe low-frequency gain and the peaking response of equalizer 120. Inother embodiments a single control signal can control multipleequalization parameters, including e.g. the low-frequency gain and thepeaking response,

Four simplified frequency-response diagrams 165, 170, 175, and 180 inthe lower portion of FIG. 1 depict the approximate effects of adjustingthe low-frequency and high-frequency gain of equalizer 120 in oneembodiment. As shown in diagram 165, increasing the value of signalLFadj tends to increase the gain of equalizer 120 at low frequencies.With reference to diagram 170, increasing the value of signal HFadjtends to decrease the peak response of equalizer 120 around a particular(high) frequency of interest. Diagram 175 shows how the broadbandfrequency response of equalizer 120 is adjusted by moving signals LFadjand HFadj together in opposite directions. Diagram 180 shows how theequalization frequency response of equalizer 120 is adjusted by movingsignals LFadj and HFadj together in the same direction. Equalizer 120can equalize incoming signals by attenuating or amplifying somefrequency components more than others, or by a combination ofamplification and attenuation.

The LFadj signal from adaptation engine 135 adjusts the low-frequencygain of equalizer 120. The HFadj signal from adaptation engine 135,adjusts the peaking response of equalizer 120. Signals LFadj and HFadjare combinations of the α[1:0] signals that indicate the broadband gain(AGCadj) and equalization emphasis (EQadj) desired. The remainingadjustment signals α[N:2] are measures of the remaining ISI attributesdue to the prior data symbols stored within buffer 160.

FIG. 2 depicts adaptation engine 135 in accordance with one embodiment.Adaptation engine 135 includes a series of synchronous storage elements200 and tap-value generators 205 that together generate, from data anderror samples Data_(n) and Err_(n), tap values α[1:0] for equalizer 120and α[N:2] for DFE 125. The data and error samples are received onrespective input ports, while the α values are conveyed to equalizer 120and DFE 125 via the corresponding adaptation-engine output ports.Tap-value generators 205 each compare incoming error signals Err_(n)with either a current data sample Data_(n) or one of N−1 prior datasamples to compute tap values α[N:0]. Element 210 shows the arithmeticlogic utilized to generate LFadj and HFadj signals from AGCadj and EQadj(α[1:0]). Increasing the value of signal HFadj decreases the peakingresponse of equalizer 120 in this embodiment.

FIG. 3 details an embodiment of a tap-value generator 205 of FIG. 2 thatgenerates a tap value using a sign-sign, least-mean-squared (LMS)algorithm. Generator 205 includes an XOR gate 300, logic 302 to convertthe unsigned XOR output to a signed number, a multiplier 305 to scalethe signed number by a constant μ, an adder 310, and a register 315. XORgate 300 compares the corresponding data and error samples and presentsits output to multiplier 305 via converter 302. The data and errorsamples represent the signs of the sampled values, so XOR gate 300 andconverter 302 collectively have the effect of multiplying the signs andpresenting the result to multiplier 305. Multiplier 305 multiplies theresulting product by a selected gain step size μ for the filter tap.Adder 310 adds the output from multiplier 305 to the current contents ofregister 315, which is then updated with the new count. Register 315thus accumulates a count representative of the α value for the filtertap associated with the data samples of a particular latency. The αvalue for the filter tap is, in turn, representative of the ISIcontribution of that filter tap to the present symbol. Ideally, each αvalue exactly offsets the respective ISI contribution. Perfection isdifficult to obtain in practice, however, and the optimal tap valuestend to vary with e.g. temperature and supply-voltage. Tap valuegenerator 205 thus adaptively maintains representative α values thatapproximate the respective ISI contributions.

FIGS. 4A-4D are waveform diagrams illustrating how tap-value generator205 generates the values for taps α₀ (AGCadj) and α₁ (EQadj) inaccordance with one embodiment. Turning first to FIG. 4A, a signal trace400 represents an incoming analog signal Veq′ over two symbol timest_(n-1) (the window for prior data Data_(n-1)) and t_(n) (the window forcurrent data Data_(n)), in a case where signal conveys a data value of 1at each symbol time. In this embodiment, Vr is equal to zero. Broadbandgain adjustments are based upon the current sampled data value Data_(n)and the current sampled error value Err_(n). The sampled error is notshown; however, it can be seen that error sample Err_(n) for FIG. 4Awould be zero because the value of trace 400 is less than Dlev in thetime interval for t_(n). In that case, the AGCadj is incremented toincrease the broadband gain of equalizer 120. The same holds true forthe example of FIG. 4C. In FIGS. 4B and 4D, however, the current valueof Veq′ is greater than Dlev, indicating that the sign of Err_(n) isone, in which case tap value AGCadj is decremented to reduce thebroadband gain.

Returning to FIG. 4A, adjustments to EQadj are based upon the priorsampled data value D_(n-1) and the current sampled error value Err_(n).As noted previously, error sample Err_(n) for FIG. 4A is zero becausethe value of trace 400 is less than Dlev in the current time interval.Also evident in FIG. 4A, the value Veq′ for the prior sample timet_(n-1) is positive (i.e., D_(n-1)=1) because Veq′ is greater thanreference voltage Vr (e.g., zero volts). In that case, the EQadj isincremented to simultaneously decrease the high-frequency and increasethe low-frequency gain of equalizer 120. The high-frequency tap valueEQadj is likewise incremented if the current error signal is a one andthe prior data signal is a zero, as shown in FIG. 4D. On the other hand,EQadj is decremented, to simultaneously increase the high-frequency anddecrease the low-frequency gain, if the current error sample has thesame value as the prior data sample, conditions that are represented inFIGS. 4B and 4C.

The forgoing error comparisons are based upon the upper signal leveldefined by voltage Dlev and applied via amplifier 140. Adaptation engine135 only updates the tap values α[N:0] based upon measurements that takeplace when the current data sample Data_(n) is a logic one. Adaptationengine 135 therefore includes a data filter, not shown, to preventupdates when the current sample Data_(n) is a logic zero. Otherembodiments can include a second amplifier/sampler pair to generateerror samples, such as by comparing the incoming signal Veq′ with thelower data level —Dlev, or the reference voltage to amplifier 140 can bevaried over a number of values or ranges of values to facilitateadditional testing and error-correction methods.

FIG. 5 depicts three eye diagrams 500, 505, and 510 that illustrate theimpact on an incoming signal Veq′ of adjusting signals AGCadj and EQadj.Beginning with diagram 500, a signal eye 515 is of relatively lowamplitude with respect to a desired data level Dlev. In this case, usingthe method described above in connection with FIGS. 4A-4D, the broadbandgain of equalizer 120 may be increased to expand eye 515. With referenceto diagram 505, the gain would continue to increase stepwise until eye515 expanded such that signal level Dlev was in the center of the upper“fuzz” band 520. At the center of the fuzz band, the error sample(Err_(n)) from sampling logic 130 would exhibit an equal likelihood ofsampling a one or a zero when the current data D_(n)=1, thus there wouldbe no further net change in AGCadj.

We next consider the impact of adjusting value EQadj. Assuming DFE 125is doing a reasonable job of cancelling the ISI associated with thepost-cursor values for taps two through N, the remaining ISI at Veq′contributing to the width of fuzz band 520 is assumed to be largely aresult of first post-cursor ISI. Using the method described above inconnection with FIGS. 4A-4D, the equalizer gain of equalizer 120 wouldbe increased or decreased as necessary to reduce the amplitude of fuzzband 520. The adjustment would continue stepwise until eye fuzz band 520diminished in the manner depicted in diagram 510 of FIG. 5. Thereafterthe EQadj, the al tap, would experience an equal likelihood ofincrementing and decrementing.

FIG. 6 schematically depicts equalizer 120 of FIG. 1 in accordance withone embodiment. Equalizer 120 includes two nearly identical stages 600and 605, the second of which is depicted as a black box for ease ofillustration. Other embodiments include more or fewer stages, or othercircuit topologies with similar frequency responses. Equalizer stage 600includes a pair of differential input transistors 615 and 620 withrespective loads 625 and 630. Source degeneration is provided by aresistor 635, a transistor 640, and a pair of variable capacitors 645and 650. The capacitance provided by transistors 645 and 650 is inparallel with resistor 635 and transistor 640 from a differentialsmall-signal perspective, so the net impedance between the sources oftransistors 615 and 620 decreases with frequency. As a consequence, thegain of equalizer stage 600 increases with frequency. The resistancethrough transistor 640 can be adjusted to change the source-degenerationresistance, and thus to alter the low-frequency response of stage 600.The capacitance through capacitors 645 and 650 can be selected to alterthe peaking response (high frequency gain) of stage 600.

In an alternative embodiment, source degeneration is provided by one ormore metal-insulator-metal (MIM) capacitors connected in parallel withresistor 635. The MIM capacitors can be used instead of or in additionto capacitors 645 and 650. Other control mechanisms might also be usedto alter the source-degeneration resistance, as by digitally switchingin different sizes and combinations of resistors. In still otherembodiments the DC gain adjustment is supported via a separategain-control amplifier, or is omitted altogether.

A DAC 655 converts the digital equalization setting LFadj[3:0] from e.g.adaptation engine 135 of FIG. 1 to a gate voltage for transistor 640.The value of the equalization setting thus determines the resistancebetween the sources of transistors 615 and 620, and consequently the lowfrequency gain of equalizer stage 600. In one embodiment, the outputvoltage from DAC 655 increases as setting LFadj[3:0] increases from 0000to 1111. This maximum output represents the lowest resistance betweenthe sources of transistors 615 and 620, and consequently the highestgain setting for stage 600. The output voltage of a similar DAC (notshown) in stage 605 performs a similar function as DAC 655 in stage 600.

FIG. 7 details an embodiment of variable capacitor 645 of FIG. 6:capacitor 650 is identical. Capacitor 645 includes a number ofcapacitor-connected transistors 700 and respective select transistors705 controlled by signal HFadj. The areas, and thus the capacitances, oftransistors 700 can vary from one to the next (e.g., their areas can bebinary coded) for added granularity, or can be thermometer coded toreduce adjustment glitches that might otherwise occur when switchingbetween values. Increasing values of HFadj[3:0] represent decreasingamounts of capacitance in the degeneration network, and thereforedecreasing high-frequency gain.

FIG. 8 schematically depicts a bias-voltage generator 800 for use withequalizer 120 of FIG. 6. A resistor 805 and transistors 810 and 815 forma half-circuit replica of equalizer stage 600, with the inputcommon-mode voltage Vin_com applied to the gate of transistor 810. Afeedback loop including an amplifier 820 and a pair of transistors 825and 830 sets the voltage on the inverting (−) terminal of amplifier 820equal to the voltage applied to the non-inverting (+) terminal. In anembodiment in which supply voltage Vdd is 1.2 volts, a resistor dividerprovides one-volt to the non-inverting terminal of amplifier 820. Theresulting bias voltage Vbias to stages 600 and 605 then establishes aone-volt common-mode output voltage for those stages. In someembodiments, lower common-mode voltages are avoided to ensure thattransistors 615 and 620 of FIG. 6 are always in saturation.

In the foregoing description and in the accompanying drawings, specificterminology and drawing symbols are set forth to provide a thoroughunderstanding of the present invention. In some instances, theterminology and symbols may imply specific details that are not requiredto practice the invention. For example, the interconnection betweencircuit elements or circuit blocks may be shown or described asmulti-conductor or single conductor signal lines. Each of themulti-conductor signal lines may alternatively be single-conductorsignal lines, and each of the single-conductor signal lines mayalternatively be multi-conductor signal lines. Signals and signalingpaths shown or described as being single-ended may also be differential,and vice-versa. Similarly, signals described or depicted as havingactive-high or active-low logic levels may have opposite logic levels inalternative embodiments.

A signal driving circuit is said to “output” a signal to a signalreceiving circuit when the signal driving circuit asserts (orde-asserts, if explicitly stated or indicated by context) the signal ona signal line coupled between the signal driving and signal receivingcircuits. The output (input) of a signal driving (receiving) circuit isgenerically referred to as an output (input) port. Circuit elements arecontrolled by application of control signals to respective controlports.

An output of a process for designing an integrated circuit, or a portionof an integrated circuit, comprising one or more of the circuitsdescribed herein may be a computer-readable medium such as, for example,a magnetic tape or an optical or magnetic disk. The computer-readablemedium may be encoded with data structures or other informationdescribing circuitry that may be physically instantiated as anintegrated circuit or portion of an integrated circuit. Although variousformats may be used for such encoding, these data structures arecommonly written in Caltech Intermediate Format (CIF), Calma GDS IIStream Format (GDSII), or Electronic Design Interchange Format (EDIF).Those of skill in the art of integrated circuit design can develop suchdata structures from schematic diagrams of the type detailed above andthe corresponding descriptions and encode the data structures oncomputer readable medium. Those of skill in the art of integratedcircuit fabrication can use such encoded data to fabricate integratedcircuits comprising one or more of the circuits described herein.

While the present invention has been described in connection withspecific embodiments, variations of these embodiments will be obvious tothose of ordinary skill in the art. For example, the depictedembodiments are signal-data-rate (SDR) systems, but other embodimentsmay support e.g. double-data-rate (DDR) or quad-data-rate (QDR)operation instead of or in addition to SDR operation. Furthermore, thereceivers described above employ current-mode signaling, but might alsobe adapted to employ voltage-mode schemes in which signals are conveyedas modulated voltages. Voltage thresholds may also be employed in thelatter case by simply converting current signals to voltage forcomparison with a voltage reference. In addition, embodiments of theinvention may be adapted for use with multi-pulse-amplitude-modulated(multi-PAM) signals, and PrDFE taps can be inserted after equalizer 120.Moreover, some components are shown directly connected to one anotherwhile others are shown connected via intermediate components. In eachinstance the method of interconnection, or “coupling,” establishes somedesired electrical communication between two or more circuit nodes,terminals, or ports. Such coupling may often be accomplished using anumber of circuit configurations, as will be understood by those ofskill in the art. Therefore, the spirit and scope of the appended claimsshould not be limited to the foregoing description. Where U.S. lawapplies, only those claims specifically reciting “means for” or “stepfor” should be construed in the manner required under the sixthparagraph of 35 U.S.C. § 112.

What is claimed is:
 1. An integrated circuit comprising: acontinuous-time equalizer including an input port to receive an inputsignal, an output port, and a first control port to control equalizationparameters of the continuous-time equalizer; a decision-feedbackequalizer coupled to the output port of the continuous-time equalizer,the decision-feedback equalizer including a second output port and asecond control port to control equalization parameters of thedecision-feedback equalizer; a first sampler to sample a differencebetween the second output port of the decision-feedback equalizer and afirst reference signal to produce first samples; a second sampler tosample a difference between the second output port of thedecision-feedback equalizer and a second reference signal to producesecond samples; and an adaptation engine coupled to the first samplerand the second sampler to receive the first samples and the secondsamples, the adaptation engine to issue control signals to the firstcontrol port of the continuous-time equalizer and to the second controlport of the decision-feedback equalizer responsive to the first samplesand the second samples.
 2. The integrated circuit of claim 1, thecontinuous-time equalizer exhibiting a low-frequency gain and a peakingresponse.
 3. The integrated circuit of claim 2, the adaptation engine toadjust the low-frequency gain responsive to comparisons between thefirst samples and the second samples.
 4. The integrated circuit of claim3, the adaptation engine to adjust the low-frequency gain in dependenceon the first samples and corresponding ones of the second samples. 5.The integrated circuit of claim 1, the input signal expressing a seriesof symbols, the continuous-time equalizer to reduce intersymbolinterference associated only with a most-recently received symbol in theseries of symbols.
 6. The integrated circuit of claim 5, thedecision-feedback equalizer to reduce the intersymbol interference onlyfrom symbols other than the most-recently received symbol in the seriesof symbols.
 7. The integrated circuit of claim 1, wherein thecontinuous-time equalizer exhibits at least one of frequency-selectiveamplification and frequency-selective attenuation.
 8. The integratedcircuit of claim 1, wherein the adaptation engine generates, from thefirst samples and the second samples, a tap value representative offirst post-cursor inter-symbol interference in the input signal, andwherein the adaptation engine applies the tap value to the first controlport.
 9. The integrated circuit of claim 8, wherein the adaptationengine generates, from the first samples and the second samples, asecond tap value representative of second post-cursor inter-symbolinterference for the input signal, and wherein the adaptation engineapplies the second tap value to the second control port.
 10. Theintegrated circuit of claim 8, wherein the tap value represents a DClevel of the input signal.
 11. A method comprising: applyingcontinuous-time equalization to an input signal to produce a firstequalized continuous-time signal; applying decision-feedbackequalization to the first equalized continuous-time signal, the applieddecision-feedback equalization to provide a second equalizedcontinuous-time signal; sampling the second equalized continuous-timesignal relative to a first reference to produce first samples; samplingthe second equalized continuous-time signal relative to a secondreference to produce second samples; controlling the continuous-timeequalization responsive to the first samples and the second samples; andcontrolling the decision-feedback equalization responsive to the firstsamples and the second samples.
 12. The method of claim 11, furthercomprising generating measures of DC attenuation of the input signal.13. The method of claim 12, wherein the measures of DC attenuation aregenerated from the first samples and the second samples.
 14. The methodof claim 12, wherein the controlling the continuous-time equalizationcomprises adjusting the continuous-time equalization responsive to themeasures of DC attenuation.
 15. The method of claim 11, the input signalexpressing a series of symbols, the continuous-time equalizationreducing intersymbol interference associated only with a most-recentlyreceived symbol in the series of symbols.
 16. The method of claim 15,the decision-feedback equalization reducing the intersymbol interferenceonly from symbols other than the most-recently received symbol in theseries of symbols.
 17. The method of claim 11, the continuous-timeequalization providing at least one of frequency-selective amplificationand frequency-selective attenuation.
 18. The method of claim 11, furthercomprising generating, from the first samples and the second samples, atap value representative of first post-cursor inter-symbol interferencein the input signal, and applying the tap value to the continuous-timeequalization.
 19. The method of claim 18, further comprising generating,from the first samples and the second samples, a second tap valuerepresentative of second post-cursor inter-symbol interference for theinput signal, and applying the second tap value to the decision-feedbackequalization.
 20. The method of claim 18, wherein the tap valuerepresents a DC level of the input signal.